Level shifter and method of calibration

ABSTRACT

A level shifter includes a signal generator that generates differential signals on a first output and a second output. A first capacitor is coupled between the first output and a first node and a second capacitor is coupled between the second output and a second node. A third capacitor is coupled between the first node and a first voltage potential, wherein the capacitance of the third capacitor is variable. A fourth capacitor is coupled between the second node and the first voltage potential, wherein the capacitance of the fourth capacitor is variable.

CROSS-REFERENCE TO RELATED APPLICATION

This application claims priority to U.S. Provisional Patent ApplicationSer. No. 62/315,471, filed Mar. 30, 2016, entitled HIGH CMTI CAPACITIVETERMINATED LEVEL SHIFT ARCHITECTURE, naming Nathan Richard Schemm asinventor, which is hereby fully incorporated herein by reference for allpurposes.

BACKGROUND

Voltage translators or level shifters are devices that resolve mixedvoltage incompatibility between different parts of a system that operatein multiple voltage domains. They are common in many complex electronicsystems, especially when interfacing with legacy devices. With theadvent of wide-bandgap semiconductors, the switching speeds of levelshifters are increasing. However, present level shifters do not have therequired high common-mode transient immunity (CMTI) with propagationtimes that are fast enough to handle these high switching speeds.

SUMMARY

A level shifter includes a signal generator that generates differentialsignals on a first output and a second output. A first capacitor iscoupled between the first output and a first node and a second capacitoris coupled between the second output and a second node. A thirdcapacitor is coupled between the first node and a first voltagepotential, wherein the capacitance of the third capacitor is variable. Afourth capacitor is coupled between the second node and the firstvoltage potential, wherein the capacitance of the fourth capacitor isvariable.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic diagram of a portion of a switching power supply.

FIG. 2 is a schematic diagram of an example of a level shifter of thepower supply of FIG. 1 that is tunable so as to increase common modetransient immunity.

FIG. 3 is an example of a signal generated by the pulse generator ofFIG. 2 in response to an input voltage.

FIG. 4 is an example of a signal at the output of an amplifier of FIG. 2in response to a pulse generated by the pulse generator of FIG. 2.

FIG. 5 is a detailed schematic diagram of an example of the firstdifferential amplifier of FIG. 2.

FIG. 6 is a flow diagram describing a method of calibrating a levelshifter, such as the level shifter of FIG. 2

DETAILED DESCRIPTION

Level shifters with high common-mode transient immunity (CMTI) and lowpropagation delay are disclosed herein. The high CMTI enables the levelshifters to operate at high switching frequencies in applications suchas driving high voltage field-effect transistors (FETs). In someexamples, the level shifters drive high-side signal translations for FETdrivers of wide-bandgap power FETs in high voltage switching powersupplies. Such wide-bandgap FETs can include gallium nitride and siliconcarbide (GaN and SiC) power FETs. With the emergence of suchwide-bandgap semiconductors, switching speeds of switching powersupplies are increasing, which is creating greater demands on the gatedrivers and level shifters within the switching power supplies.Traditional switching power supplies reduce switching losses byimplementing wide-bandgap drivers having slew-rates that are higher thancurrent level shifters can support without errors.

FIG. 1 is a schematic diagram of a portion of a switching power supply100. The power supply 100 includes a controller 104 that is coupled to aswitching portion 106, whereby the controller 104 drives a FET Q11 and aFET Q12 in the switching portion 106. The FET Q11 is sometimes referredto as a high-side FET and the FET Q12 is sometimes referred to as alow-side FET. In some examples, the FETs Q11 and Q12 are wide-bandgapGaN FETs with drain/source breakdown voltages of approximately 600V. TheFETs Q11 and 012 are examples of switches that may be implemented in theswitching portion 106. Other switching devices may be implemented in thepower supply 100 as known by those skilled in the art. The power supply100 enables a high voltage swing between a transmitter (not shown) and areceiver (not shown).

The drain of FET Q11 is coupled to a voltage source V11, which is a highvoltage source and in some examples the voltage source V11 has a voltagepotential between zero and 600V. The source of FET Q12 is coupled to avoltage potential, which in the example of FIG. 1, is a ground.

The controller 104 includes control circuitry 110 that may receive andoutput a plurality of signals and voltages to drive the switchingportion 106. In the example controller 104, the control circuitry 110receives a control signal at a node N11. In some examples, the controlssignals include a pulse width modulated (PWM) signal, which controls orsets the timing of the switching portion 106. In other examples, thecontrol circuitry 110 may have other inputs coupled thereto. The controlcircuitry 110 has an output 112 coupled to the input of a level shifter120 and an output 124 coupled to the input of a driver 126 that drivesthe FET Q12.

The level shifter 120 enables the controller 104 to operate the FET Q11at a high voltage when the controller 104 itself is operated at a muchlower voltage. The level shifter 120 has an output 130 that is coupledto a driver or amplifier 132, which controls the gate voltage of the FETQ11. Likewise, the driver 126 controls the gate voltage of the FET Q12.The driver 126 operates at a voltage VDD, such as 5V, relative to avoltage VSS, which may be ground. The level shifter 120 and the driver132 may operate at a small voltage, but their ground reference V_(HS)may be much higher than the VSS potential. Accordingly, the voltagedifference between the ground reference V_(HS) and a supply voltageV_(HB) may be VDD or 5V.

When the FET Q12 turns off, the FET Q11 turns on and the voltage V_(HS)rapidly slews up to the voltage V11. The output of the level shifter 120also slews up with the voltage V_(HS), which produces a very fastcommon-mode transient for the level shifter 120. High speed switchingpower supplies require a driver with very good common-mode transientimmunity (CMTI) to withstand the high slew rates of wide-bandgap devicessuch as the FETs Q11 and Q12. Many switching power supplies furtherrequire low propagation time and propagation matching to support highswitching frequencies. Furthermore, many switching power suppliesrequire level shifters with low quiescent current consumption. Levelshifters are disclosed herein that have high CMTI, operate at highswitching frequencies, and draw low quiescent current.

FIG. 2 is a schematic diagram of an example of a level shifter 200 thatis tunable to increase CMTI. The level shifter 200 is coupled to aninput that may be coupled to the node N11 of FIG. 1. The input 202 iscoupled to a pulse generator 206 that converts the input signal at theinput 202 to a plurality of differential pulses that are output on nodesQ and Q′. The signal output on the node Q is referred to as the signalV21 and the signal output on the node Q′ is referred to as V22. In otherexamples, signal generation devices other than the pulse generator 206may be implemented to generate differential signals representative ofthe input signal on node N11.

The nodes Q and Q′ are coupled to a plurality of drivers 208. The lastof the drivers 208 are a driver 210 and a driver 212 that are coupled toor powered by a variable voltage source 216. The variable voltage source216 sets the amplitude of the signals V23 and V24 at the output of thedrivers 210 and 212. As described in greater detail below, the variablevoltage source 216 varies the amplitudes of the signals V23 and V24 tocalibrate the output amplitude of the level shifter 200. In someexamples, the plurality of drivers 208 are implemented with a singledriver coupled to the Q node and a single driver coupled to the Q′ node.

A capacitor C21 is coupled between the driver 210 and a node N21 and acapacitor C22 is coupled between the driver 212 and a node N22. Thecapacitors C21 and C22 isolate the voltage potential V_(HS) from lowvoltage circuitry, such as the drivers 208 and the pulse generator 206.A capacitor C23 is coupled between the node N21 and a voltagetermination V_(T). A capacitor C24 is coupled between the node N22 andthe voltage termination V_(T). The voltage termination V_(T) may be aplurality of different voltages as described herein. The capacitors C23and 024 are variable or able to be trimmed to improve the CMTI at nodesN21 and N22 as described in greater detail below. In some examples, thecapacitance values of the capacitors C23 and C24 are greater than thecapacitance values of the capacitors C21 and C22. The capacitors C21 andC23 form a voltage divider at node N21 and capacitors C22 and C24 form avoltage divider at node N22. The signals V23 and V24 are typically highfrequency signals or contain high frequency components, such as stepfunctions, which are able to pass through capacitors C21 and C22 andbecome a differential signal at nodes N21 and N22. Common-mode signalsare generated on N21 and N22 in response to CMTI across the levelshifter 200. During calibration, the ratio of C21 to C23 is closelymatched to the ratio of C22 to C24 to minimize the differential outputproduced on nodes N21 and N22 in response to CMTI. If the ratios are notclosely matched, transient common mode voltages may cause delays and/orerrors in processing of the signals V23 and V24 as described herein.

Differential inputs of a differential amplifier 220 are coupled to thenodes N21 and N22. The differential amplifier 220 processes the signalsV21 and V22 as described herein. Differential inputs of anotherdifferential amplifier 222 are also coupled to the nodes N21 and N22.The differential amplifier 222 measures the differential transientresponse on the nodes N21 and N22 during a transient test and generatesa signal V_(TEST), which is proportional to the differential transientresponse. The signal V_(TEST) is input to a processor 224 that trims thecapacitance values of the capacitors C23 and C24 in response to thesignal V_(TEST).

A resistor R21 couples a voltage source V_(CM) to the node N21 by way ofa switch SW21 and a resistor R22 couples the voltage source V_(CM) tothe node N22 by way of the switch SW21. The state of the switch SW21 isset by the processor 224 and the switch SW21 serves to charge the nodesN21 and N22 to the voltage V_(CM), which is the common mode voltage ofthe differential amplifier 220. The charges on the nodes N21 and N22 areanalyzed by the processor 224 to determine the proper capacitance valuesof the capacitors C23 and C24 to maximize CMTI as described herein.

In the example of FIG. 2, the output of the differential amplifier 220is coupled to the input of a second differential amplifier 230. In theexample of FIG. 2, the differential amplifier 220 has a very goodhigh-frequency common mode rejection ratio (CMRR). For example, a twovolt swing over a two nanosecond period may produce a maximum 2 mVdifferential swing on the output of the differential amplifier 220. TheCMRR of the differential amplifier 220 is a factor that limits the CMTIof the level shifter 200. The differential amplifier 220 is sometimesreferred to herein as the first stage. Common-mode voltage swings on thenodes N21 and N22 have little effect on the gain of the differentialamplifier 220. The differential amplifier 230 has moderate gain, whichmay be less than the gain of the differential amplifier 220.Furthermore, the differential amplifier 230 has low output impedance todrive large loads of components coupled to the outputs of thedifferential amplifier 230 as described herein.

The differential output of the differential amplifier 230 is coupled toa first RC network, which in turn is coupled to the inputs of acomparator 234. The differential output of the differential amplifier230 is also coupled to a second RC network, which in turn is coupled tothe inputs of a comparator 236. A high output of the differentialamplifier 230 is coupled to capacitors C25 and C26 and a low output ofthe differential amplifier 230 is coupled to capacitors C27 and C28. Thecapacitors C25 and C27 are coupled to inputs of the comparator 234 andcapacitors C26 and C28 are coupled to inputs of the comparator 236.Resistors R23 and R24 couple the inputs of the comparator 234 to avoltage source V25 and resistors R25 and R26 coupled the inputs of thecomparator 236 to a voltage source V26. The voltage source V25 sets athreshold for triggering voltage transitions on the output of thecomparator 234 and the voltage source V26 sets a threshold fortriggering voltage transitions on the output of the comparator 236. Theoutputs of the comparators 234 and 236 are coupled to the input of alatch 240 that, in the example of FIG. 2, includes two NAND gates. Theoutput of the latch 240 is coupled to the gate of transistor Q11. Insome examples, an amplifier or driver (not shown) is coupled between thelatch 240 and the gate of transistor Q11.

FIG. 3 is an example of the signal V21, FIG. 2, generated by the pulsegenerator 206 in response to the signal received on node N11. The signalV22 is the complement of the signal V21. The signal V21 shown in FIG. 3is an example of a plurality of different signal types that may begenerated by the pulse generator 206. In the example of FIG. 3, thepulse generator 206 generates either positive or negative pulses on therising and falling edges of the input signal at node N11. The pulsegenerator 206 further generates pulses to keep the level shifter 200active. The input signal has a rising edge 300, which causes the pulsegenerator 206 to generate a pulse 302 that has a predetermined pulsewidth t31. In the example of FIG. 3, the predetermined pulse width t31is 3 ns. The pulse 302 is referenced by the letter M to denote that itis a main pulse generated at the beginning of a transition in the inputsignal. Insurance pulses, referenced as the letter I, are transmittedafter a predetermined time t32 from the main pulses. In the example ofFIG. 3, an insurance pulse 306 is shown being transmitted after apredetermined time t32 from the main pulse 302. In the example of FIG.3, the predetermined time t32 between the main pulse and the insurancepulse is 20 ns. If the input signal has not transitioned after apredetermined time t33, the pulse generator 206 generates a keep pulse,referenced by the letter K. In the example of FIG. 3, the pulsegenerator 206 has generated a keep pulse 310 at a time t33 from thegeneration of the insurance pulse 312.

The pulses in the signals V23 and V24 conduct through the capacitors C21and C22, respectively, and are terminated at the capacitors C23 and C24,which may have capacitance values substantially larger than thecapacitance values of the capacitors C21 and C22. The differences incapacitance values form capacitive voltage dividers between the outputsof the drivers 210, 212 and the nodes N21, N22. In the examplesdescribed herein, the voltage dividers have a large ratio, such as330V/V. The ratio is chosen such that the full voltage swing of theinput relative to the output is equal to at least half of the overallcommon-mode range of the differential amplifier 220.

As described above, the capacitors C23 and C24 are trimmable in order totrim out the common-mode to differential conversion which wouldotherwise occur due to mismatched ratios in the capacitance values ofC21/C23 and C22/C24 as described herein. Trimming the capacitors C23 andC24 may be performed after assembly of the level shifter 200, such asduring testing. The input signal on node N11 is inactive during testing,so the pulse generator 206 does not generate any pulses. The processor224 closes switch SW21, which charges the capacitors C21, C22, C23, andC24 by way of the common mode voltage V_(CM). A high impedance situationis then created by the processor 224 opening switch SW21, which allowsany differential errors on the nodes N21 and N22 to be held there forreadout through the amplifier 222. The V_(HS) voltage is then swept to ahigh voltage relative to the input of the level shifter. Then, anydifferential errors related to capacitor mismatch are held on thecapacitors C21, C22, C23, and C24 and read by the processor 224 via thedifferential amplifier 222. If the ratio of the capacitance values ofthe capacitors C21 to C23 is equal to the ratio of the capacitancevalues of the capacitors C22 to C24, then the voltage on node N21 willbe equal to the voltage on node N22. The amplifier 222 measures thedifference between the voltages on nodes N21 and N22 and outputs thedifference to the processor 224. In the example described herein, theamplifier 222 has a gain of twenty, but other gain values may beimplemented as required by specific applications. The processor 224 thendetermines the values of the capacitors C23 and C24. It is noted that insome examples, the processor 224 is separate from the level shifter 220.

As described above, mismatch in the ratios of the capacitances of thecapacitors C21, C22, C23, and C24 creates a common-mode to differentialconversion and trimming the capacitors C23 and C24 improves thecommon-mode to differential conversion performance. The trimming processis converted into a low frequency trim by disconnecting the commonvoltage source V_(CM) from resistors R21 and R22, which sets DC voltageson the capacitors C23 and C24. The DC voltages on the capacitors C23 andC24 are the voltage on the nodes N21 and N22, respectively. Then, thecommon-mode is swept and any errors created by the mismatch are left onthe capacitors C23 and C24 and are measured via the amplifier 222.Sweeping the common mode includes moving the high-voltage side of thelevel shifter 200 from 0V where it was when the switch SW21 was open toa high voltage. The high voltage develops across the C21 and C22. Themeasuring may be accomplished over a long period due to a slow timeconstant associated with the capacitors C23 and C24. The amplifier 222can be double-sampled to eliminate any offset error in the amplifieritself. For example, the output of the amplifier 222 may be sampledbefore SW21 is opened and both inputs are still at the same voltagepotential, and then sampled again after the error on N21 and N22 havesettled. The difference of the two readings gives an error which isindependent of the offset of the amplifier 222.

As described above, the output signal or voltage of the amplifier 222 isreceived by the processor 224. The processor 224 then analyzes thevoltage output by the amplifier 222 to determine which of the capacitorsC23 and/or C24 needs to be trimmed and how much trimming needs to occurso the above-described ratios are equal. The process of measuring thecommon-mode to differential conversion may be repeated after an initialtrimming to be sure that the capacitors C23 and C24 have been trimmedcorrectly.

FIG. 4 is a graph showing an example signal 400 at the output of theamplifier 230 in response to a pulse generated by the pulse generator206, FIG. 2. The graph shows a noise margin between a positivecomparison threshold and a negative comparison threshold where thesignal 400 is not detectable. As shown in FIG. 4, a CMTI induced signalis present in the signal 400, but it is within the noise margin and willnot induce errors. The signal 400 exceeds the positive comparisonthreshold and enters a signal margin at a time 402. The signal amplitudeof the signal 400 determines how far in excess of the noise margin thesignal 400 extends. If the signal amplitude is too low, the signal 400will not be detected above the noise margin.

The level shifter 200 provides the ability to set the threshold level ofthe comparators 234 and 236 to achieve a signal, such as the signal 400of FIG. 4 with appropriate signal and noise margins. In the examplesdescribed herein, the signal amplitude is set to twice that of the noisemargin. The process includes adjusting the output of the drivers 210 and212 to lower voltages. In the example of FIG. 2, the output voltages ofthe drivers are set to half of their normal operating voltage by way ofthe variable voltage source 216 supplying a lower or half voltage to thedrivers 210 and 212. The voltages V25 and V26 are then adjusted to wherethe signal 400 just exceeds the noise margin. The output of thecomparators 234, 236 or the output of the latch 240 may be monitored todetermine if the signal 400 has exceeded the noise margin. The processor224 then instructs the variable voltage source 216 to output the fullvoltage to the drivers 210 and 212, which returns the output of thedrivers 210 and 212 to their full voltages. The signal amplitude 400 isthen as shown in FIG. 4.

FIG. 5 is a schematic diagram of an example of the differentialamplifier 220 of FIG. 2. The first stage of the amplifier 220 providesbenefits that improve the operation of the level shifter 200, FIG. 2.The capacitors C23 and C24 may be terminated with a voltage VDD, ground,or a voltage in between ground and VDD. The amplifier 220 has a veryhigh common-mode rejection ratio (CMRR), which is achieved by takingadvantage of the inputs and nodes N21 and N22, which can be loaded withhigh capacitance without affecting the circuit amplifier 220.

FIG. 6 is a flow diagram describing a method of calibrating a levelshifter, such as the level shifter 200 of FIG. 2. The method commencesat step 600 with coupling a first node to a first voltage potential. Thefirst node is coupled to a first capacitor that is coupled to a signalgenerator, a second capacitor coupled to a second voltage potential, anda first input to a first differential amplifier. Step 602 includescoupling a second node to the first voltage potential. The second nodeis coupled to a third capacitor that is coupled to the signal generator,a fourth capacitor coupled to the second voltage potential, and a secondinput to the first differential amplifier. Step 604 includes decouplingthe first voltage from the first node and the second node. Step 606includes sweeping a voltage across the level shifter to generate adifferential voltage between the first node and the second node. Step608 includes measuring the voltage difference between the first node andthe second node. Step 610 includes adjusting the capacitance value of atleast one of the second capacitor and the fourth capacitor in responseto the measuring.

Although illustrative embodiments have been shown and described by wayof example, a wide range of alternative embodiments is possible withinthe scope of the foregoing disclosure.

What is claimed is:
 1. A circuit comprising: a signal transmitter havinga first transmitter terminal and a second transmitter terminal; a signalreceiver having a first receiver terminal and a second receiverterminal; a first capacitor coupled between the first transmitterterminal and the first receiver terminal, and having a firstcapacitance; a second capacitor coupled between the second transmitterterminal and the second receiver terminal, and having a secondcapacitance; a third capacitor coupled between the first receiverterminal and a terminal voltage source, and having a third capacitancedefining a first ratio over the first capacitance; and a fourthcapacitor coupled between the second receiver terminal and the terminalvoltage source, and having a fourth capacitance defining a second ratioover the second capacitance, the second ratio matching the first ratio.2. The circuit of claim 1, further comprising: a common mode voltagesource; a first switch coupled between the first receiver terminal andthe common mode voltage source; and a second switch coupled between thesecond receiver terminal and the common mode voltage source.
 3. Thecircuit of claim 1, further comprising: a differential amplifier havinga first input coupled to the first receiver terminal, and a second inputcoupled to the second receiver terminal.
 4. The circuit of claim 3,wherein: the third capacitance and the fourth capacitance are eachadjustable; the differential amplifier is configured to generate a testsignal based on a voltage difference between the first and secondreceiver terminals when the first and second receiver terminals aredecoupled from a common mode voltage source; and at least one of thethird capacitance or the fourth capacitance is adjusted based on thetest signal such that the second ratio matches the first ratio.
 5. Thecircuit of claim 4, further comprising: a processor coupled to receivethe test signal from the differential amplifier, and configured toprovide instructions to adjust at least one of the third capacitance orthe fourth capacitance.
 6. The circuit of claim 5, wherein the processoris configured to decouple the common mode voltage source from the firstand second receiver terminals while the test signal is being generated.7. The circuit of claim 5, wherein the processor is configured todetermine a common mode transient immunity between the first receiverterminal and the second first receiver terminal based on the testsignal.
 8. The circuit of claim 1, wherein the signal transmitterincludes: a first buffer configured to transmit a first signal to thefirst transmitter terminal; a second buffer configured to transmit asecond signal to the second transmitter terminal; and a variable voltagesource coupled to the first and second buffers.
 9. The circuit of claim1, wherein the first capacitance is different from the secondcapacitance.
 10. A method of calibrating a level shifter, the methodcomprising: coupling a first node to a first voltage potential, thefirst node being coupled to a first capacitor coupled to a signalgenerator, a second capacitor coupled to a second voltage potential, anda first input to a first differential amplifier; coupling a second nodeto the first voltage potential, the second node being coupled to a thirdcapacitor coupled to the signal generator, a fourth capacitor coupled tothe second voltage potential, and a second input to the firstdifferential amplifier; decoupling the first voltage from the first nodeand the second node; sweeping a voltage across the level shifter togenerate a differential voltage between the first node and the secondnode; measuring the voltage difference between the first node and thesecond node; and adjusting the capacitance value of at least one of thesecond capacitor and the fourth capacitor in response to the measuring.11. The method of claim 10, wherein inputs of a second differentialamplifier are coupled to the first node and the second node, and whereinthe measuring comprises measuring the output voltage of the seconddifferential amplifier.
 12. The method of claim 10, wherein theadjusting comprises adjusting the capacitance value of at least one ofthe second capacitor and the fourth capacitor to equalize the voltagesbetween the first node and the second node.
 13. The method of claim 12,wherein equalizing the voltages on the first node and the second nodecomprises making the voltage on the first node within a predeterminedvalue of the voltage on the second node.
 14. The method of claim 10,wherein the first voltage potential is a function of the common-moderejection ratio of the first differential amplifier.
 15. The method ofclaim 10, further wherein adjusting the capacitance value of at leastone of the second capacitor and the fourth capacitor in response to themeasuring indicating that the voltage difference between the first nodeand the second node exceeds a predetermined value.
 16. A methodcomprising: generating, by a transmitter, a first signal having a firstvoltage amplitude during a calibration mode; generating, by thetransmitter, a second signal having a second voltage amplitude during anoperation mode, the second voltage amplitude greater than the firstvoltage amplitude; and setting a threshold voltage of a comparator in areceiver to detect the first signal during the calibration mode and thesecond signal during the operation mode.
 17. The method of claim 16,wherein the first voltage amplitude is half of the second voltageamplitude.
 18. The method of claim 16, wherein the first and secondsignals each comprises a pair of differential signals.
 19. The method ofclaim 16, further comprising: transmitting the first and second signalsfrom the transmitter to the receive via a capacitor.
 20. The method ofclaim 16, wherein the second voltage amplitude is greater than the firstvoltage amplitude by a noise margin.